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Broadband electronically tunable IC active radiating elements and power combiners.


Considerable effort has been directed toward the development of microwave and mm-wave hybrid and monolithic integrated circuits (ICs). Recently active devices have been integrated directly with planar antennas to create active radiating elements. By designing the antenna and oscillator on a single substrate, transition/transmission line losses are avoided and complete monolithic integration becomes a step closer. Merits of this technology become more apparent at mm- and sub-mm-wavelengths. At these frequencies, solid-state devices produce very low power levels while waveguide dimensions and tolerance become difficult to maintain. The use of active antennas and spatial or quasi-optical power combining techniques can overcome these limitations.|1,2~ General power combining techniques have been reviewed previously.|1~ Power combining results and the current state of quasi-optical power combining technology also has been described previously.|3~

In simple terms, active antennas are those antennas directly integrated with active devices to generate RF power. Active antennas have been realized using two-terminal and three-terminal devices. FETs and other three-terminal solid-state devices have shown higher DC-to-RF conversion efficiencies than two-terminal devices. However, diodes reach much higher operating frequencies. The active antenna provides a tank circuit, matching transformer or feedback mechanism to sustain oscillations. A group of active antennas can be injection-locked for phase-coherency and power combining.

The microstrip patch antenna has been integrated with diodes and transistors for active, planar, low cost radiating elements.|4-8~ The microstrip structure provides a resonant patch for oscillations and a ground plane for efficient heat sinking. However, as an active antenna element, the patch has exhibited very narrow tuning ranges with high cross-polarization levels and wide output power deviations. Recently, an active rectangular patch with symmetrical Gunn diodes obtained higher output power and lower cross-polarization levels.|9~ At mm-wavelengths, the patch radiator dimensions become increasingly small and sensitive to thickness variations. Although varactors have been integrated in waveguide and microstrip Gunn oscillators,|10-12~ other solid-state devices are not easily integrated on the patch structure for tuning purposes. A tuning diode and bipolar transistor have recently been integrated within a multilayer patch radiator to obtain 4.4 percent tuning of the operating frequency.|13~ An alternative active antenna to the microstrip patch radiator is the uniplanar notch.

The notch antenna has many desirable characteristics, including broad impedance matching bandwidth, uniplanar construction, good reproducibility, scalability to sub-mm-wavelengths and ease of integration of solid-state devices. The length of the notch can be increased to create a traveling-wave antenna, similar to the linear-tapered slot antenna.|14~ The flare of the notch antenna is another important design parameter, as seen with the exponentially-tapered Vivaldi antenna.|15~ Important work has been reported on the impedance and radiation characteristics.|16-18~ The design flexibility of this type of antenna makes it useful for many diverse applications. Its scalability is useful well into the sub-mm-wave region.

As operating frequencies increase, solid-state devices produce lower power and active antennas and spatial power techniques become increasingly important in meeting the needs of most applications. Power-combining techniques include chip-level, circuit-level and spatial combiners. Spatial and open resonators involve a large number of transistors or diodes. The individual free-running oscillators must be injection-locked to produce a coherent higher power RF source. Injection locking may be obtained via mutual coupling, external feedback or an external source. The result of using spatial or quasi-optical power combining techniques is to create a single, coherent and high power signal from many low power radiating sources. Spatial or quasi-optical power combining is not limited by size, ohmic or dielectric losses, or moding problems and allows the combination of a greater number of active devices. The patch radiator and grid has been the element of choice for most active antennas and in power combining.|19-23~ As an end-fire radiator with wide bandwidth and scalability to sub-mm-wave-lengths, the notch antenna may serve as a valuable element for power combining when patch radiator antenna dimensions become impractically small. Monolithic integration of a Gunn diode has been demonstrated on a uniplanar resonator|24~ and its results are directly applicable to the active notch design. The described circuit, a varactor, tunes the notch electronically. However, another device, either a Gunn and/or IMPATT, can be inserted for increased power.

Previously, a FET has been integrated with a notch antenna|25~ using a slotline and coplanar waveguide (CPW) to create a 20 GHz receiver front-end. Recently, wideband active notch antennas integrated with Gunn and varactor diodes were developed.|26,27~ Two circuit configurations were devised. The first circuit configuration was a bias-tuned Gunn oscillator in a CPW resonator using a stepped-notch antenna. The notch is coupled to the center of a CPW resonator that is physically perpendicular to the slotline. The second circuit configuration was a varactor-tuned Gunn oscillator in a CPW resonator using a smooth-tapered notch antenna. The notch is coupled to one end of a CPW resonator that is physically parallel to the slotline.

The bias-tuned Gunn oscillator configuration exhibits a clean, stable bias-tuned signal from 9.2 to 9.47 GHz with a power output of 14.2 |+ or -~ 1.5 dBm. Bias tuning creates a wide deviation in power output and the resonator configuration introduces a strong cross-polarization component. A varactor introduced into the oscillating circuit provides a clean, stable signal with constant power output over a wide electronic tuning bandwidth. The varactor-tuned Gunn oscillator configuration exhibits a tuning bandwidth from 8.9 to 10.2 GHz with an output power of 14.5 |+ or -~ 0.8 dBm. The spectral purity and tuning range are comparable to results achieved in waveguide and microstrip oscillators. The theoretical tuning curve obtained from a circuit model agrees well with experimental results.

Injection-locking experiments show a locking gain of 30 dB with a locking bandwidth of 30 MHz at 10.2 GHz. Power combining experiments of two varactor-tuned active notch antenna elements, 8 mm apart in a broadside configuration, have achieved over 70 percent combining efficiency throughout the tuning range with a maximum of 129.2 percent at 9.7 GHz. A varactor-tunable active antenna is attractive for power combining in electronically agile radar and communication systems.

To demonstrate this technology for mm-wavelengths, a 37 GHz active notch antenna was built using a Gunn diode on a 0.762 mm RT/duroid 5880 substrate. Over 15 mW output power was achieved with a good radiation pattern. Further integration of another Gunn or IMPATT diode would increase the active element's output power. Integration of a varactor would show a similar tuning range as exhibited by the X-band notch.

The Active CPW Stepped-Notch Antenna

Figure 1 shows the active CPW stepped-notch antenna configuration. The circuit consists of a stepped-notch antenna coupled to a CPW resonator via a slotline. A Gunn diode is placed in a heat-sink at the open terminals of the resonator. The notch is formed by many step transformers that match the slotline impedance to free space.

The resonator is the essential design element for improved oscillations and stability. CPW was chosen for the resonator because of its uniplanar construction and ease of integration with active devices. The CPW slots are 0.3 mm wide with a 3.5 mm separation. This arrangement provides a 50 |omega~ characteristic impedance and mates well with the 3.5 mm cap of the Gunn diode package. The length of the resonator is 0.5 |lambda~ at 10 GHz. A DC block was incorporated at the shorted end for biasing. The circuit is mounted on an L-shaped metal block that serves as a heat-sink and holder.

Sections of cascading slotlines act as impedance transformers from the coupling point to free space in the stepped-notch design. The low dielectric constant of 2.3 allows for efficient antenna radiation. The input impedance of the notch radiator was matched to the resonator at the coupling point and optimized using a transmission line equivalent circuit model. The transformer sections' lengths were designed to obtain minimum return loss throughout X-band. The circuit was fabricated on a 60 mil thick, RT/duroid 5870 substrate. To test the passive circuit, an SMA connector was soldered on the CPW resonator and the measurements were performed on a network analyzer. The stepped-notch antenna gain was measured at 9.3 and 9.6 GHz and found to be 7.1 and 7.7 dBi, respectively. Integration of the active devices slightly changes the antenna performance and introduces degenerate modes at the oscillating frequency. The gain measurements were used to calculate the oscillator power output with the Friis transmission equation

|Mathematical Expression Omitted~


|P.sub.r~ = power received |P.sub.t~ = power transmitted from the active notch antenna |lambda~ = wavelength of operation R = antenna range length |G.sub.ot~ = transmit antenna's gain |G.sub.or~ = receive antenna's gain

A Gunn device was integrated with the stepped-notch antenna. The Gunn diode produces 72 mW of RF power in an optimized waveguide circuit. The bias voltage vs. frequency and power output is shown in Figure 2. The 3 dB bias-tuning bandwidth was 275 MHz centered at 9.33 GHz with a maximum power output of 37.5 mW at 9.328 GHz.

The Varactor-Tunable CPW Active Notch Antenna

Figure 3a shows the varactor-tunable CPW active notch antenna configuration. The circuit consists of a notch antenna integrated with a varactor-tuned CPW resonator. The notch antenna couples to the resonator via a slotline. A Gunn and a varactor diode are placed on either side of a CPW resonator. The arrangement provides strong coupling between the Gunn diode and the varactor diode for increased tuning bandwidth.

The inherent DC blocks of this CPW resonator configuration allow easy integration of multiple DC-biased devices. The CPW center conductor serves as DC ground, while each of the fins of the unilateral notch serve as biasing pads for two solid-state devices. The length of the resonator is 0.5 |lambda~. The antenna's input impedance was matched at the coupling point with the resonator and the flare of the notch antenna was modified from the stepped-notch to a smooth taper for improved bandwidth performance.

A theoretical model was developed to facilitate the design. The equivalent circuit and a photograph are shown in Figures 3b and 4, respectively. Table 1 lists the circuit's specifications. The modeling of the CPW junction is based on previous work.|28~ Neglecting the junction discontinuity effects, the transformer ratio n is set equal to 1. The flare of the notch antenna was approximated by 24 sections of slotlines of different widths and impedances. The equivalent circuits of the Gunn and varactor diodes used for calculations are shown in Figure 5. The diode parasitics for the Gunn and varactor are given by the vendor. The following conditions were used to determine the oscillating frequencies at various varactor bias levels. For condition 1

/Re(|Z.sub.diode~)/|is greater than or equal to~/Re(|Z.sub.circuit~)/ (2)

and for condition 2

Im(|Z.sub.diode~) + Im(|Z.sub.circuit~)/ = 0 (3)

where Re(|Z.sub.diode~) is assumed to be -8 ||omega~.sup.29~ and the oscillations occur when condition 1 is satisfied. The oscillating frequencies are found from condition 2.
Z1 = 160                    L1 = 0.685
Z2 = 160                    L2 = 3.529
Z3 =  50                    L3 = 3.500
Z4 = 110                    L4 = 7.200
Z5 = 122                    L5 = 2.540

The dimensions of the circuit board are 1 X 2". The circuit was etched on a 60 mil thick RT/duroid 5870 substrate. To test the passive circuit, a coaxial connection was soldered onto the notch and measured on a network analyzer. The measured SWR was |is less than~ 2 throughout the X-band range of 8 to 12.4 GHz. The passive notch was then tested in an anechoic chamber for the radiation field patterns and relative gain. The relative gain |G.sub.o~ of the passive notch antenna measured at 9, 9.6 and 10.2 GHz was 8, 8.2 and 9 dBi, respectively. The gain was later used in Equation 1 to determine the active VCO notch output power.

A Gunn and varactor diode were integrated into the CPW resonator and coupled to the notch antenna via a slotline. The Gunn diode produces 80 mW of RF power in an optimized waveguide circuit while the varactor is rated for 1.6 pF at 0 V. Theoretical results were calculated using the varactor tunable notch antenna circuit model and Equations 2 and 3. Figure 6 shows the experimental results and the theoretical tuning curve obtained for varying varactor bias levels from 0 to 30 V. Considering the large deviation of active device modeling parameters, the theoretical model agrees well with the experimental results. A frequency tuning range of 8.9 to 10.2 GHz was achieved for varactor voltages of 0 to 30 V. There are no mode jumps and the signal spectrum remains clean and stable over the 14 percent electronic tuning bandwidth. The VCO power output is 14.5 |+ or -~ 0.8 dBm. The spectrum of the received signal from the varactor-tunable notch antenna is comparable to the spectrum of the active stepped-notch. The E- and H-field patterns and the cross-polarization patterns for the varactor-tunable active notch antenna at 10.2 GHz are shown in Figure 7. The back radiation remains 10 dB below the maximum.

Injection-Locking and Power-Combining Experiments

Injection-locking experiments with an external sweep oscillator source were performed to determine the locking-gain and locking-bandwidth of the active VCO notch antenna configuration. The test measurement set-up is shown in Figure 8. Equation 1 determined the injection-lock signal power from the transmitted power |P.sub.t~ and the free-running oscillator power was determined from the received power |P.sub.r~. Injection-locking experiments were performed throughout the electronic tuning range. The locking-gain vs. locking bandwidth results are shown in Figure 9. They are comparable to previous patch radiator antenna experiments. A locking gain of 30 dB with a locking bandwidth of 30 MHz was obtained at 10.2 GHz. The circuit's Q-factor is found to be 21.5 and can be calculated using|30~

|Q.sub.e~ = 2|F.sub.0~/|delta~F |square root of |P.sub.i~/|P.sub.o~~ (4)

The locking gain is defined as

|G.sub.L~ = 10 log (|P.sub.o~/|P.sub.i~) (dB) (5)


|Q.sub.e~ = external Q-factor |F.sub.o~ = operating frequency |delta~F = injection-locking bandwidth |P.sub.i~ = injection-lock signal power |P.sub.o~ = free-running oscillator power

Quasi-optical combiners using Fabry-Perot resonators and spatial power combiners have the potential of combining many solid-state devices at mm-wave frequencies. To demonstrate the feasibility of a spatial power combiner, two notch antennas were set up 8 mm apart in a broadside array configuration. To achieve efficient power-combining, the active notch antenna elements must injection-lock each other through mutual coupling. Power-combining experiments of two injection-locked, active VCO notch antennas were conducted throughout the electronic tuning range at 100 MHz increments. The power combining efficiency |eta~ is defined by

|Mathematical Expression Omitted~

|P.sub.1~ = power of active notch 1 use |G.sub.ot~ = |G.sub.o~ |G.sub.o~ = antenna gain of a single passive notch antenna |P.sub.2~ = power of active notch 2 use |G.sub.ot~ = |G.sub.o~ |P.sub.combiner~ = injection-locked, power-combined signal's power use |G.sub.ot~ = 2|G.sub.o~

All power calculations are computed from Equation 1. For the |P.sub.1~ and |P.sub.2~ calculations, the gain of a single passive antenna was used for the transmitting active antenna gain |G.sub.ot~. For the |P.sub.combiner~ calculations, the array gain was assumed to be twice the gain of a single passive antenna. The power combining efficiencies measured at 9.4, 9.7 and 10 GHz were 90, 129.2 and 75 percent, respectively, as shown in Figure 10. The combining efficiency of over 100 percent at certain frequencies can be attributed to improved impedance matching in two mutually-coupled oscillators as compared to a single nonoptimized oscillator. Similar results have been reported by another power combiner.|31~ The H-plane field pattern and cross-polarization measurements at 9.6 GHz for the power combiner active VCO notch are shown in Figure 11. The 3 dB beamwidth of the array was 53|degrees~, compared to 78|degrees~ for a single element.

This particular notch element can also include two diodes, either a Gunn and/or IMPATT, for increased element power. The notch array can be injection-locked via mutual coupling, dielectric feedback or an external low power source. The notch array can be used as a transmitter with an external injection locking source for mechanical beam steering. The array can also feed a reflector or quasi-optical resonator. Figure 12 shows two power combining configurations.

A Ka-Band Active Notch Antenna

To demonstrate the mm-wave application of this technology, a Ka-band active notch antenna was fabricated on a 0.762 mm thick RT/duroid 5880 substrate. A Gunn diode was integrated into the CPW resonator in the same way it was integrated into the stepped-notch antenna. Figure 13 shows the received spectrum from the active Ka-band notch antenna. Equation 1 was used to calculate a 15 mW oscillator output power at 37 GHz. Figure 14 shows a photograph of the Ka-band notch antenna. Integration of another solid-state device can serve for either increased power, switching or voltage tunability.


The Gunn device has been integrated with a stepped-notch antenna using a CPW resonator. The configuration was modified to incorporate a varactor diode and form a varactor-tuned active antenna element. Over 14 percent electronic tuning range was achieved with a constant power output. Two of these active VCO antenna elements were injection-locked to each other via mutual coupling over a 1.1 GHz bandwidth from 9 to 10.1 GHz. Power combining was demonstrated using mutual coupling over this range with over 70 percent combining efficiency. The Ka-band notch antenna demonstrated scalability to the mm-wave range.

These circuits offer simple, lightweight, low cost, reproducible and uniplanar active wideband tunable sources for many microwave applications. The configuration allows insertion of multiple devices for increased output of a single element. Using these elements in planar arrays with injection-locking and power combining techniques will enable higher power levels. The wide varactor tuning range is useful for frequency modulated communication links, radar and electronic warfare applications. The circuits are amenable to monolithic circuit integration for mass production and should have many applications in frequency-agile transmitters at microwave and mm-wave frequencies.


This work was supported in part by the US Army Research Office and Texas' Higher Board of Education Advanced Technology Program. The authors thank the Rogers Corporation for the substrate material and acknowledge Yon-Hui Shu's suggestions and technical assistance which led to the completion of this investigation.


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Julio A. Navarro received his BSEE and his MSEE from Texas A&M University in 1988 and 1990, respectively. From 1985 to 1991, he worked for General Dynamics, where he was a member of the avionic systems design, advanced technology and system engineering, emitters and intelligence, antenna systems and radar cross section research groups. He has served as a teaching and research assistant at Texas A&M. As a research assistant, he has designed varactor tunable endfire radiating elements, switchable and tunable uniplanar filters and Gunn VCOs. He has also developed Ka-band aperture-coupled circular patch antennas for a NASA-Lewis-Texas Instruments project. His current research interest include high-Q uniplanar resonators and quasi-optical power combiners. Navarro holds an NSF engineering fellowship and is currently completing his PhD.

Kai Chang received his BSEE degree from the National Taiwan University, Taipei, Taiwan; his MS degree from the State University of New York, Stony Brook; and his PhD from the University of Michigan, Ann Arbor, in 1970, 1972 and 1976, respectively. From 1972 to 1976, he worked as a research assistant at the University of Michigan's Cooley Electronics Laboratory in the Microwave Solid-State Circuits Group. From 1976 to 1978, he was employed by Shared Applications Inc. From 1978 to 1981, he worked for Hughes Aircraft Co.'s Electron Dynamics Division. From 1981 to 1985, he worked for TRW Electronics and Defense. In 1985, Chang joined the electrical engineering department of Texas A&M University as an associate professor, and in 1988, he was promoted to professor. In 1990, he was appointed E-Systems Endowed Professor of electrical engineering. His current interests are in microwave and mm-wave devices and circuits, microwave ICs, microwave optical interactions and antennas. Chang received the Special Achievement Award from TRW in 1984, the Halliburton Professor Award in 1988, the Distinguished Teaching Award in 1989, and the Distinguished Research Award in 1992 from the Texas A&M University. Chang is a fellow of the IEEE.
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Title Annotation:integrated circuits
Author:Navarro, Julio A.; Kai Chang
Publication:Microwave Journal
Date:Oct 1, 1992
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