A substrate integrated waveguide bandpass filter using novel defected ground structure shape.
Substrate integrated waveguide (SIW), early introduced by Wu et al., has attracted plenty of attention and utilized in different applications due to its merits of higher quality factor, easy integration, low cost, high power handling and mass production [1-5]. The half-mode substrate integrated waveguide (HMSIW) has been developed for microwave and millimeter wave applications [6, 7]. The HMSIW gave good performance and almost 50% reduction in SIW size. However, compared with the microstrip circuitry, the conventional rectangular cavity resonator of the HMSIW is still too large to make its physical size be small enough.
In order to reduce the size of SIW components, several technologies have been proposed. The first one is to utilize the multilayer low temperature co-fired ceramic (LTCC) and printed circuit board (PCB) process to develop the three dimensional substrate integrated circular and elliptic cavity (3D SICC and SIEC) [8-10] and the folded substrate integrated waveguide [11, 12]. It can make the SIW cavities to be folded and to couple with each other in the vertical direction, so that the component's size can be reduced greatly. However, size-reducing in this method is at the cost of fabrication complexity and cost. The second method to do size-miniaturization is to use metamaterial resonators, such as the complementary split ring resonator (CSRR) , the complementary sprial resonator (CSR)  and the composite right/left-handed structures (CRLH) , which exhibit negative permittivity and permeability. As they are utilized in the SIW, a passband propagation below its cut-off frequency of the dominant mode can be achieved. However, it is quite unacceptable in microwave and millimeter wave applications that the measured insertion losses of all experimental filters in [13,14] are not low enough. Another promising approach is the defected ground structure (DGS), which is etched as featured patterns in the ground plane of planar circuit board to change the ground current distribution and generate resonance, so as to increase stopband reduction and reduce size of the circuit [16-19]. However, since all DGS referred above are etched into the bottom side of the circuit boards, it is inconvenient to integrate them with other components in one metallic cavity. Solution for this drawback is to etch DGS in the plane of signal transmission lines of the coplanar waveguide (CPW) [20, 21] and SIW .
2. ANALYSIS OF SIW-DGS CELL
The proposed SIW-DGS cell is shown in Figure 1. Since the DGS is etched into the top metal cover of SIW, it is quite convenient to do system integration. For this proposed SIW-DGS cell, its bandpass function is the composite high-low (Hi-Lo) type, i.e., it is a combination of the highpass guided wave function of SIW and the bandgap function of DGS.
As discussed in [1,2], the characteristic cutoff frequency of the SIW is mainly controlled by its width a in Figure 1. A SIW with a and h, the height of the substrate, can be equivalent to a conventional rectangular waveguide with the same height, h, and width, [a.sub.EFF_SIW], which is shown as
[a.sub.EFF_SIW] = a - [[(2R).sup.2]/0.95D], (1)
where R and D are the vias' radius and the spacing between two adjacent vias, respectively. As shown in , the Equation (1) is valid for D < [[lambda].sub.0] [square root of ([[epsilon].sub.r]/2)] and D < 8R within [+ or -] 5% precision, where A0 and [[epsilon].sub.r] are the wavelength in free space and the relative permittivity of the substrate, respectively. As R increases, the small error will appear, which has been discussed in . To improve the precision, a more accurate width aEFF_srw can be rewritten as
[a'.sub.EFF_SIW] = a - 1.08 [[(2R).sup.2]/D] + 0.1 [[(2R).sup.2]/a], (2)
where good precision can be obtained under conditions of D/(2R) < 3 and a/R > 5, which has been reported in . To make the effect of D to be more intuitive, some simulated results of one SIW with different D is given in Figure 2, where the relative permittivity [[epsilon].sub.r] = 2.2, a = 7.0 mm, h = 0.508 mm, and R = 0.25 mm. According to Figure 2(a), as D increases, although the cutoff frequency almost has no change, the rejection under the cutoff frequency improve greatly, which is a positive influence. However, the negative influence is the return loss in the passband becomes worse with the increasing of D. Furthermore, as shown in Figure 2(b), the insertion loss in the passband increases from 0.12dB to over 0.16dB as D increases from 0.8 mm to 1.2 mm. This may be due to the fact that energy leakage increase with increasing D/(2R).
In this SIW-DGS cell, the upper cutoff frequency is mainly controlled by the DGS. For proposed DGS shape, size of its slots and the spacing between adjacent slots are very important parameters to affect the bandgap point and the bandgap strength. To make the characteristics of proposed DGS clear, these parameters are discussed in details, with the same conditions as [[epsilon].sub.r] = 2.2, a = 7.0 mm, h = 0.508 mm, R = 0.25 mm, and D = 1mm. The influence of the outer slots' size is simulated and shown in Figure 3. It can be found that as c or d increases, i.e., size of the outer slots gets larger, the upper cutoff frequency and bandgap point shifts lower, while the lower cutoff frequency nearly keeps the same. That means the larger the size of the DGS is, the lower the bandgap frequency is. On one hand, for the same d, the longer c is, the stronger the bandgap strength is. Inversely, for the same c, as d gets longer, the bandgap strength gets weaker. For different group choosing of c and d with "2c+d = 7.0mm", such as "c = 1.25 mm, d = 4.5 mm", "c = 1.00 mm, d = 5.0 mm" and "c = 0.75 mm, d = 5.5 mm", it can be seen that the longer c is, the lower the upper cutoff frequency and the bandgap point are, and the stronger the bandgap strength is. As shown in Figure 3, the bandgap strength of "c = 1.25mm, d = 4.5mm" is over 60dB, which is much stronger than those of "c = 1.00 mm, d = 5.0 mm" and "c = 0.75 mm, d = 5.5 mm". In a components design using this SIW-DGS cell, the lengths of c and d should be tradeoff and tuned to meet the design requirement.
For the inner slow-wave slots, their sizes, mainly determined by the transverse length c2 and longitudinal length d2, have little influence on the passband transmission characteristics, but can affect the upper stopband performance heavily. According to Figure 4, for the same d2, as c2 gets longer, the transmission characteristics nearly have no change in the passband, and only have a very small difference in the bandgap point. But for the same c2, as d2 gets longer, the upper stopband transmission behavior gets worse rapidly and heavily. As shown in Figure 4, when d2 increases from 0.3 mm to 0.6 mm, lots of spurs occur in the stopband, which is unacceptable in filter design. So when this cell is used in the design, the longitudinal length of the slow-wave slots should be tuned to reach a good stopband spurious performance.
Figure 5 shows the simulated results of the proposed SIW-DGS cell with different spacing of the two outer slots, i.e., b. It can be seen that as b gets wider, the upper cutoff frequency of proposed cell shifts lower, but the bandgap point shifts higher, which means that the upper skirt selectivity gets worse and that the bandwidth gets narrower. The bandgap strength of "b = 2.0 mm" is much stronger than those of "b = 1.0 mm" and "b = 3.0 mm". As "b = 3.0 mm", spur occurs in the stopband. So it can be predicted that there is an optimal value of " b" around 1.0 mm to 2.0 mm, which can help to optimize the performance in components design.
3. FILTER DESIGN
Based on the proposed SIW-DGS cell, a X-band wideband bandpass filter is designed. In this filter design, two elements are pivotal: one is the identical property of the SIW-DGS cell, which has been presented exhaustively in Section 2, and the other is the cascade of two adjacent cells. In the cascade of two cells, "s", width of the metal spacing between them, is the decisive factor to affect the transmission characteristics, such as the upper skirt selectivity, the spur's location and its strength. As shown in Figure 6, the wider the spacing is, the worse the skirt selectivity is, the lower the spur occurs and the stronger its strength is.
The configuration of proposed filter is shown in Figure 7. It consists of three cascaded cells in the middle and two CPW-SIW transitions at input/output (I/O) ports in order to achieve high external quality factor. Its optimized dimensions and electric field distribution are demonstrated in Table 1 and Figure 8, respectively. According to its electric field distribution, most of the energy is restrained in the DGS slots, so that most of the transmission can be carried out around the slots, which can reduce the transmission loss resulted from the coupling aperture in conventional cavity filter.
4. EXPERIMENTAL RESULTS
Using a single layer PCB process, the filter, as shown in Figure 9, is fabricated on a RT/Duroid 5880 substrate with relative permittivity of 2.2 [+ or -] 0.02, loss tangent 0.001 (at 10 GHz) and a thickness of 0.508 mm, and is measured by the Agilent Vector Network Analyzer N5245A. Its simulated and measured S-parameters are compared in Figure 10. From the measured results, the filter has a central frequency of 9 GHz, a fractional bandwidth of 32% and return loss better than 20 dB in the whole passband. The measured maximum insertion loss is 0.81 dB, about 0.15 dB worse than that of the simulated one, and the ripple in the passband is less than 0.2 dB. As shown in Figure 10, the measured upper 3-dB cutoff frequency is a bit lower than that of the simulated one, which might be caused by the variation of substrate's permittivity and the inaccuracy in fabrication. In Figure 11, whatever for measured and simulated results, both variations of the group delay of [S.sub.21] are less than 0.2 ns, which is quite smooth for a microwave filter. So, the proposed filter can be used as a linear phase filter in broadband wireless communication and digital microwave system.
To inspect the performance of proposed filter clearer, some comparisons between the proposed filter and several previous SIW cavity filters, folded SIW filter and SIW-DGS filters reported in the references are summarized in Table 2. According to the comparisons, although the return losses of the filters in [24,25] are quite good, their insertion losses are much higher than that of the proposed one, which is mainly contributed by the transmission loss of inter-coupling between adjacent cavities and the radiation loss. Compared with the proposed filter, due to the narrow band, the cross-layer aperture coupling and multilayered structure, the multilayered filters reported in [9,10,12,27] have higher insertion loss, larger sizes and higher fabrication price. Although the SIW-DGS filter in -2 and the SIW-CSRR filter in  have good performance on the whole, as a tradeoff of insertion loss, return loss and size, they are inconvenient to do system integration owing to their DGS and CSRR etched into the bottom metal cover. Although the SIW-CPW filter in -3 is with compact size, low insertion loss and is easy to do system integration, its return loss is not so good as well as the in-band ripple of its insertion loss is over 0.5 dB, which is much larger than that of the proposed one. This larger ripple will make the error vector magnitude (EVM) of the signal to be much worse when such kind of filter is used in the digital microwave system with high-order quadrature amplitude modulation (QAM). It is obvious that the proposed filter can achieve lower insertion loss, better return loss and more compact size on the whole. Comparison of the group delay of [S.sub.21] is not available here because most of the filters presented in references haven't been reported.
A X-band wideband bandpass filter based on the novel SIW-DGS cell is designed, fabricated, and measured. The measured results are in good agreement with the simulated one in group delay, in-band ripple and linear-like phase. Compared with some reported filters operating at similar frequency, the proposed filter has better overall performance in insertion loss, return loss and size.
This work was supported in part by Greating Technology (Beijing) Co. Ltd. and in part by Foundation of State Key Laboratory of Millimeter Wave, Southeast University, under Grant No. K201111.
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Yongmao Huang1 (,) Zhenhai Shao (1,2), *, and Lianfu Liu (1)
(1) School of Communication and Information Engineering, University of Electronics Science and Technology of China, Chengdu 611731, China
(2) State Key Laboratory of Millimeter Waves, Southeast University, Nanjing 210096, China
* Corresponding author: Zhenhai Shao (email@example.com).
Received 4 November 2012, Accepted 14 December 2012, Scheduled 19 December 2012
Table 1. Dimensions of proposed filter with three cascaded SIW-DGS cells. Symbol Quality (mm) Symbol Quality (mm) ws 1.52 61 2 w 0.5 c1 1 a 14.8 d1 4.65 wi 0.3 t1 0.52 Lt 1.1 62 2.1 D 1 c2 0.95 R 0.25 d2 4.7 bi 0.6 t2 0.49 we 2.6 63 2 x2 3.8 c3 1.05 di 0.25 d3 4.6 ci 0.3 t3 1.8 Table 2. Comparison with filters presented in references. Reference Order Topology Central frequency filter and fractional bandwidth  3 HMSIW- Slot 7.8 GHz and 22%  2 3D SICC 10.05GHz and 3.3%  2 3D SICC/SIEC 10.03 GHz and 3.4%  4 HMSIFW 9.9725 GHz and 0.8%  3 SIW-CSRR 8.15 GHz and 23%  3 SIW cavity 10.09 GHz and 3%  3 CSRR cavity 5.03 GHz and 6.4% -1 3 SIW-EBG 13GHz and 46.1% -2 3 SIW-DGS (bottom) 11.75 GHz and 55.32% -3 3 SIW-CPW 11.75 GHz and 55.32%  3 SIFW 10 GHz and 1%  3 SIW-CSRR (bottom) 9.4 GHz and 30%  4 7.8 GHz and 6% Folded SIW Proposed 3 SIW-DGS 9 GHz and 32% Reference Insertion Return Size filter loss (dB) loss (dB) ([[lambda].sup.2.sub.0])  1.5 > 10 > 0.20  1.9 > 18.5 > 0.54  1.65 > 19.5 > 0.53  1.5 > 12 > 0.47  2.16 11.6 0.35  3.13 > 13.5 1.49  3.9 > 19 0.58 -1 2.5 ~ 5 > 10 0.39 -2 > 1.4 > 11 > 0.26 -3 > 0.8 > 11 > 0.25  2.7 > 20 0.65  1 16 0.47  2.62 > 10 0.15 Proposed < 0.81 > 20 0.36
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|Author:||Huang, Yongmao; Shao, Zhenhai; Liu, Lianfu|
|Publication:||Progress In Electromagnetics Research|
|Date:||Mar 1, 2013|
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