A single-balanced mixer with a coplanar balun.
As the lower microwave frequencies are becoming more popular for civilian applications, such as cable TV and mobile communications, there is a growing need for low-cost portable spectrum analyzers for installation and maintenance. Low-cost components must be designed that offer integrability for reduced size in a simple housing and tuning simplicity for efficient line production and maintenance. These requirements call for structures where sensitive components, such as the diodes of the front end mixer, and tuners are on the same side of the substrate.
A coplanar hybrid single-balanced diode mixer for upconversion of RF from 9 kHz to 2.3 GHz in the first stage of a spectrum analyzer has been designed. A constant IF of 4 GHz is maintained using LO frequencies in the range from 4 to 6.3 GHz. This mixer includes a coplanar balun and coplanar waveguide-microstrip transitions for connecting the LO and RF-IF ports to the rest of the RF system.
Isolation between LO signal and both RF and IF signals is necessary, especially for low RF. Indeed for a quasi zero RF, the IF equals the LO frequency. If a strong LO component is present at the IF port, it might saturate the next element in the signal-processing chain.
A common form of a single-balanced mixer is shown in Figure 1. The diodes are identical and the transformer is center-tapped. The LO signal is applied to the diodes as a balanced mode. The RF and IF signals (small compared to the LO signal) are applied unbalanced. The impedance Z stands for the RF-IF diplexer seen from the mixer. The principle of superposition applies here for the small RF and IF signals. Let the LO pump the diodes and the RF source be turned off (V = 0). Voltage of the primary winding is
[V.sub.p] - [V.sub.g] = [V.sub.lo] Voltages of the secondary windings are
[V.sub.a] - [V.sub.g] = [V.sub.g] - [V.sub.b] = [V.sub.lo]' Therefore, voltage of point D is
[V.sub.d] - [V.sub.g] = 0 where D ignores the LO signal. Let an RF signal now be applied to D V << [V.sub.lo]). Voltages of the secondary windings become
[V.sub.a] - [V.sub.g] = [V.sub.lo]'+ V'
[V.sub.g] - [V.sub.b] = [V.sub.lo]' - V' (V' << [V.sub.lo]') Therefore, voltage of the primary winding is unchanged and P ignores the RF and IF signals. Isolation between the LO port and RF-IF port is thus achieved.
A balun is used to transform the unbalanced signal coming from the LO to a balanced one, performing the role of the transformer in Figure 1. A planar balun is made by etching a slot in the ground plane of the planar line, which brings in the LO signal. Planar baluns have been reported previously, but they involved multilayer structures. A coplanar waveguide (CPW)-slotline transition is modified and used as a monolayer balun.
The CPW-slotline transition shown in Figure 2a has been used in the MIC design. The CPW-slotline transition can be simplified and made smaller for use as a balun if ports 1 and 2 are fed with an even mode ([a.sub.1] = [a.sub.2]), as represented by the E-field lines. The transition is then divided into two identical parts by a magnetic wall; near the wall the field lines run parallel to it. Each half behaves qualitatively similar to the transition shown in Figure 2b, where the CPW is replaced by an asymmetrical CPW (ACPW). Comparison of the E-field lines in Figures 2a and 2b shows how even mode propagation in Figure 2a is used to achieve the modified structure in Figure 2b. The size is reduced in this new transition because the ground plane of the ACPW extends over one side only. Moreover, the use of air-bridges is avoided.
The ACPW-slotline transition is used as a balun. The ACPW and slotline are terminated with an open and a short, respectively, which serve as tuners, as shown in Figure 3a. The diodes are series connected right at the transition across the slot. Zero-bias is maintained by the DC path of the slotline. A simple equivalent circuit is shown in Figure 3b. The design of the ACPW consists of a gap g of the ACPW at a fixed value; a width w computed for a characteristic impedance of 50 [omega]; a static analysis of an ACPW on an infinitely thick substrate using a conformal transform; a numerical method to account for wall proximity, thickness of dielectric and conductor; and a program for finite-element analysis of quasi-TEM transmission lines, which is modified for smaller computer storage and greater execution speed, and used to compute static line capacitances C([[epsilon].sub.r]) and C(1). Line impedance and effective permittivity are given by
[Mathematical Expression Omitted] and
[[epsilon].sub.eff] = [epsilon].sub.0] C([[epsilon].sub.r])/C(1)
The ACPW-slotline balun, shown in Figure 3a, is built for a separate reflection test, with a short-open calibration set, shown in Figure 4, which shifts the measurement reference plane R-R' to the transition. The diodes are replaced by a standard surface-mounted 50 [omega] resistor soldered across the slot.
Mixer to LO Port Connection
The transition shown in Figure 5 connects the ACPW to the microstrip from the LO. The two lines share a strip, while their respective ground planes are connected together with metal ribbons through a via-hole made in the substrate. A transition is built for a separate reflection test, along with a short-open calibration set, shown in Figure 6. In order to cancel reflections from the imperfect 50 [omega] resistor, which terminates the ACPW, some microwave absorber is laid on top of the ACPW beyond the transition.
Mixer to RF-IF Port Connection
In order to bring the RF signal to the diodes, a strip is laid in the middle of the slot, as shown in Figure 7. The slot short is shaped into an airbridge. Thus, a CPW is implemented. While carrying the unbalanced RF and IF signals as CPW even modes to be erased, the slot acts to provide a short-circuit tuning mechanism for the balun (CPW odd mode). The capacitance between the air-bridge and CPW strip is seen by the RF and IF signals as if grounded. Therefore, it does not degrade isolation. At the other end of the CPW the transition to the slot of the balun is seen by the RF and IF signals as an open circuit.
The transition shown in Figure 8 connects the CPW to the microstrip that carries RF and IF signals, but no measurement was given. It is tested in the same way as the microstrip ACPW transition. The test set of the microstrip-CPW is shown in Figure 9. The return loss of the coplanar balun, the microstrip-ACPW transition and the microstrip-CPW transition is shown in Figure 10.
A mixer is built on RT/Duroid 6010 with the thickness h = 1.27 m m, a relative permittivity [[epsilon].sub.r] = 10.2 and a conductor thickness t = 17.5 [mu]m. it should be noted that all transitions were tested on this substrate. A series pair of high barrier Schoftky diodes in an inexpensive plastic package (2.5 mm [phi]) were used. All via-hole connections in the prototype were made with copper ribbons. Series production involves metallized via-holes instead. Assembly time is minimum, as only one diode pair and one air-bridge need to be soldered on the ready-to-test integrated mixer. Dimensions of the circuit are 14 X 23 mm. A 4 X 16 X 22 mm cavity is machined in the housing bottom wall underneath the mixer. Cover height on top of the circuit is h = 5 mm. Computed width of the 50 [omega] ACPW is w = 1.4 mm for a gap g = 0.2 mm. Microstrip and CPW dimensions are computed with the Touchstone(*) program.
In the prototype, ACPW open stub and RF-IF CPW are oversized to allow tuning with various diodes. Once a particular model of diodes is selected for series production, open ACPW and air-bridge shorted slot can be shortened and the mixer made smaller. Tuning is then rarely necessary
During measurements the mixer is completely enclosed in its metal case. A layer of microwave absorbing material is bonded to the lid in order to avoid cavity resonance, which would degrade the isolation. The measurement set-up and calibration are shown in Figure 11. LO power is 10 dBm, RF input power is -20 dBm, and constant IF is 4 GHz.
Measured performance is shown in Figure 12. The conversion loss is between 5 and 6 d B for 0 < RF [less than or equal to] 2.3 GHz. Isolation between the RF-IF port and LO port is better than 29 dB for 4 [less than or equal to] [f.sub.lo] [less than or equal to] 6.3 GHz. In the RF band of interest, RF and LO port return loss is greater than 7 and 8 dB, respectively.
The mixer, shown in Figure 13, is connected to a diplexer, which is tested separately on the same kind of substrate. The diplexer is a traveling-wave filter, whose structure has the advantages of easy coupling to the RF-IF port microstrip line, good impedance matching conditions at all ports at both RF and IF and isolation between RF and IF ports. Measured transmission loss from RF-IF to IF port is 0.8 dB, and measured isolation between RF and IF port is 28 dB.
A low-cost single-balanced mixer was presented that offers manufacturing and tuning simplicity, as well as integrability with good performance. It involves four kinds of transitions between four different transmission lines in a simple design. its LO and RF-IF networks overlap for reduced space, still providing good isolation.
The author extends his thanks to his friend Stephen Herbert for his comments and suggestions on this article. (*) Touchstone is a trademark of EEsof
[1.] B.R. Hallford, "A Designer's Guide to Planar Mixer Baluns," microwaves, Dec. 1979, pp.52-57. [2.] K.C. Gupta, R. Garg and I.J. Bahl, Microstrip Lines and Slotlines, Artech House, Norwood, MA, 1979. [3.] I. Kneppo and J. Gotzman, "Basic Parameters of Nonsymmetrical Coplanar Lines," IEEE Trans. microwave Theory Tech., Vol. MTT-25, August 1977, p. 718. [4.] P.P. Silvester and R.L. Ferrari, Finite Elements for Electrical Engineers, Cambridge University Press, 1983. [5.] B. Davies, "Numerical Methods in Electromagnetics," University of Colorado, Spring 1989. [6.] M. Riaziat, S. Bandy and G. Zdasiuk, "Coplanar Waveguides for MMICs," Microwave Journal, June 1987, pp. 125-131. [7.] G. Matthaei, L. Young and E.M.T Jones, Microwave Filters, Impedance-Matching Networks and Coupling Structures, Artech House, Norwood, MA, 1980.
Denis Jaisson graduated from the Joint European Scheme, Paris, Karlsruhe, Colchester, with a degree in electrical engineering in 1985. He worked at AEG, Ulm, Germany on electronic countermeasures. He also performed research on planar electromagnetic modeling at the MIMICAD center in Colorado. Currently, he is a senior R&D engineer with GIGA-Instrumentation, France. His main interest lies in the design of planar circuits, microwave components and subsystems.
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|Date:||Jul 1, 1992|
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